In recent years, radio apparatuses are being used for data communication between apparatuses in portable electronic apparatuses, household electronic appliances, or peripheral apparatuses for personal computers. Increases in the amount of data in radio apparatuses has been accompanied by demand for increases in the transmission speed as well as energy efficiency, and UWB (Ultra-Wide Band) communication apparatuses that employ, for example, multihand OFDM (Orthogonal Frequency Division Multiplexing) are anticipated as a communication method that can meet these demands (for example, see “High Rate Ultra Wideband PHY and MAC Standard,” ECMA International Standard ECMA-368 1st Edition, December 2005, pp. 7, 14-16). There is also a demand in recent years for reducing costs in radio apparatuses by unifying communication standards to enable use in a wider range of equipment.
One known configuration that is advantageous for achieving lower costs of a radio apparatus is the direct conversion method such as shown in FIG. 1 that effects direct conversion of a radio frequency (RF) signal to baseband signal by means of a local oscillation frequency (LO) signal. FIG. 1 shows an example of a configuration of a reception apparatus of the background art that adopts the direct conversion method.
The reception apparatus shown in FIG. 1 is provided with antenna apparatus 101, low noise amplification circuit (LNA) 102, mixer 103, local oscillator (LO) 104, first variable gain amplifier (VGA) 105, low pass filter (LPF) 106, second VGA 107, and A/D converter (ADC) 108.
LNA 102 amplifies an RF signal that is received by antenna apparatus 101.
Mixer 103 mixes the RF signal supplied from LNA 102 with an LO (local oscillation frequency) signal supplied from LO 104 to produce a baseband signal.
First VGA 105 and second VGA 107 amplify the baseband signal that is supplied from mixer 103 to match the dynamic range of ADC 108.
LPF 106 eliminates the unnecessary high frequency component that is contained in the baseband signal.
ADC 108 converts the baseband signal that is composed of an analog signal to a digital signal and supplies the result to a baseband processing circuit (not shown).
The direct conversion method is capable of conversion from an RF signal to a baseband signal and from a baseband signal to an RF signal with few parts and therefore enables a radio apparatus at low cost. However, the direct conversion method is known to have the problems described below.
In the reception apparatus shown in FIG. 1, an RF signal supplied from LNA 102 and an LO signal supplied from LO 104 are mixed by mixer 103 as described above to produce a baseband signal. At this time, mixer 103 both mixes the RF signal and LO signal as well as the leakage component (hereinbelow referred to as the “LO component”) 109 of the LO signal that is leaked from LO 104 to mixer 103 by way of the substrate or power supply and the LO signal supplied from LO 104, whereby a DC component (DC offset) is generated in the baseband signal. The amount of leakage of LO component 109 to this mixer 103 is changed by the frequency of the LO signal, and the value of the DC offset that is supplied from mixer 103 therefore changes according to the frequency of the LO signal.
As described hereinabove, first VGA 105 and second VGA 107 adjust gain such that the input signal of ADC 108 is not saturated. As a result, when DC offset is contained in the output of mixer 103, first VGA 105 and second VGA 107 amplify the baseband signal by suppressing the gain such that the input signal of ADC 108 is not saturated by the DC offset. As a result, the baseband signal is not adequately amplified and the reception sensitivity drops. This phenomenon is referred to as self-mixing because LO component 109 that has leaked to the RF signal is mixed by the LO signal itself.
Regarding the causes for the appearance of DC offset in the output of mixer 103, it is known that a DC offset is also caused by, in addition to the above-described self-mixing, discrepancies in the characteristics of each part of mixer 103, but explanation here regards only DC offset produced by self-mixing.
The DC offset that appears in the offset of mixer 103 becomes an even more serious problem when an LO signal of a plurality of different frequencies is used. For example, in a UWB communication apparatus (hereinbelow referred to as simply “UWB”) that employs the above-described OFDM method, a communication band of from 3.1 GHz to 10.6 GHz is divided into 14 bands and a signal subjected to OFDM modulation is transmitted and received using a band of 528 MHz per band. In a UWB, moreover, three bands are assigned to each terminal and data are transmitted and received by switching these bands each 312.5 ns (see FIG. 2). A method of switching radio frequencies that are used as carriers with the passage of time is therefore referred to as frequency hopping.
When hopping radio frequencies in the reception apparatus of the direct conversion method shown in FIG. 1, the size of the DC offset that appears in the output of mixer 103 changes as shown in FIG. 3. At this time, when the hopping frequency component that corresponds to the hopping period approaches the band of the received signal, eliminating only the hopping frequency component becomes problematic, resulting in the blockage of a portion of the band of the received signal, deterioration of S/N, and a further reduction of reception sensitivity.
The configuration shown in FIG. 4A and FIG. 4B is known as a first example of the background art for ameliorating this problem. FIG. 4A and FIG. 4B show an example of a baseband circuit provided in a radio apparatus described in Japanese Laid-Open Patent Publication No. 2001-211098.
FIG. 4A and FIG. 4B show a configuration that detects the DC offset supplied from a VGA and controls the output voltage of the VGA such that the detected DC offset becomes 0.
The circuit shown in FIG. 4A uses an A/D converter (ADC) to convert the DC offset that appears in the output of VGA 401 to a digital signal, generates a correction signal for correcting the DC offset by means of a control circuit, uses a D/A converter (DAC) to convert the generated correction signal to an analog signal, and supplies the analog signal to the control terminal of VGA 401.
Similarly, an A/D converter (ADC) is used to convert the DC offset that appears in the output of VGA 402 to a digital signal, a correction signal for correcting the DC offset is generated by a control circuit, a D/A converter (DAC) is used to convert the generated correction signal to an analog signal, and the analog signal is supplied to a control terminal of VGA 402.
The circuit shown in FIG. 4A is further provided with switch 405 that short-circuits two signal lines for applying differential input to VGA 402. By turning ON switch 405 at the time of correcting the DC offset of VGA 402, the input signal of VGA 402 is set to zero.
The circuit shown in FIG. 4B uses an A/D converter (ADC) to convert the DC offset that appears in the output of succeeding VGA 404 to a digital signal, generates a correction signal for correcting the DC offset by means of a control circuit, and uses a D/A converter (DAC) to convert the generated correction signal to an analog signal and supplies the analog signal to the control terminal of preceding VGA 403.
The circuit shown in FIG. 4B is further provided with switches 407 in LPF 406 that is inserted between VGA 403 and VGA 404. Switches 407 are components for ameliorating the problem of the time required for correcting of the DC offset due to the delay of LPF 406, switches 407 being turned OFF at the time of correction to decrease the time constant of LPF 406 and thus shorten the convergence time required for correction.
The first example of the background art enables both the correction of the offset of a VGA and the DC offset resulting from self-mixing supplied from a mixer. The first example of the background art further enables correction of DC offset even when DC offset of different values is supplied from the mixer in synchronization with frequency hopping by executing a process similar to that described hereinabove for each hopping frequency.
The configuration described in Japanese Laid-Open Patent Publication No. 2006-203686 is a second example of the background art for correcting DC offset that appears in the output of a mixer.
As shown in FIG. 5, the second example of the background art is of a configuration provided with three sets of switch 501 and capacitor 502 connected in series and in which three high-pass filters (HPF) made up of these three sets and resistor 503 are connected to the outputs of a mixer. In the example shown in FIG. 5, three HPF are provided on each I-channel and on each Q-channel.
Switches 501 repeatedly turn ON and OFF in synchronization with frequency hopping. The operation is described by taking as an example a case in which an LO signal hops among the three frequencies F1, F2, and F3.
In the configuration shown in FIG. 5, SW#1 turns ON and SW#2 and SW#3 turn OFF when the local frequency is F1. At this time, the output signal of the mixer is supplied to the LPF by way of the HPF made up by C#1 and R, and the direct-current component of the signal that is integrated by C#1 and R is accumulated at the two ends of C#1.
When the local frequency next hops from F1 to F2, SW#1 and SW#3 turn OFF and SW#2 turns ON. At this time, the output signal of the mixer is supplied to the LPF by way of the HPF made up by C#2 and R, and a direct-current component of the signal integrated by C#2 and R is accumulated at the two ends of C#2.
When the local frequency hops from F2 to F3 as well, the direct-current component of the signal that is integrated by C#3 and R is similarly accumulated at the two ends of C#3. By repeating the above-described process, the direct-current components corresponding to each local frequency are accumulated at capacitors C#1-3 and the DC offset supplied as output from the mixer by frequency hopping is not supplied to the LPF.
The configuration described in Japanese Laid-Open Patent Publication No. 2006-020334 shown in FIG. 6 is the third example of the background art for correcting the DC offset that appears in the offset of a mixer.
The third example of the background art is a configuration in which, similar to the first example of the background art shown in FIG. 4A and FIG. 4B, the DC offset that appears in the output of an amplification unit (a VGA in the first example of the background art) is converted to a digital signal by an A/D converter (ADC) and supplied to a control unit, and the correction signal generated in the control unit is converted to an analog signal by a D/A converter (DAC) and fed back to the input of the amplification unit.
The two chief points of difference between the third example of the background art and the first example of the background art are as follows:
First, in the first example of the background art, the DC offset of a VGA is corrected by supplying a correction signal to the VGA. In contrast, the third example of the background art is a configuration in which the DC offset of an amplification unit is corrected by applying a correction signal to adder 601 that is arranged to precede the amplification unit (VGA).
Second, in the first example of the background art, the DC offset supplied from a mixer for each instance of frequency hopping is converted to an analog signal using one DAC and supplied to the VGA. In contrast, in the third example of the background art, a DAC is provided corresponding to each hopping frequency, the correction value of the DC offset that is generated at each frequency is saved in advance in a register, and the value saved in each register is converted to an analog signal by a corresponding DAC and supplied to adder 601. In a configuration that uses a plurality of DAC in this way, DAC having a comparatively slow conversion speed can be applied in a radio apparatus that requires high-speed frequency hopping.
The above-described frequency hopping is known to have an adverse effect not only upon a reception apparatus but also upon a transmission apparatus.
When the offset of a mixer and/or the DC offset of a baseband signal that is applied as input to the mixer are mixed by an LO signal in a transmission apparatus, the frequency component of the LO signal appears in the output of the mixer (this phenomenon is hereinbelow referred to as a local leak). Unwanted radiation produced by the transmission of this LO signal may result in failure to meet the standards of the radio apparatus that have been established by law.
In addition, the amount of a local leak changes according to the frequency of the LO signal that results from discrepancies in the characteristics of the parts that make up a mixer, whereby a local leak must be corrected for each frequency of the LO signal in a system that adopts frequency hopping.
The configuration described in Japanese Laid-Open Patent Publication No. 2006-238243 and shown in FIG. 7 is known as the fourth example of the background art for correcting DC offset produced in a transmission apparatus. In FIG. 7, the configuration described in Japanese Laid-Open Patent Publication No. 2006-238243 is shown simplified to an extent sufficient for grasping the essential points.
The fourth example of the background art is of a configuration in which the output of mixer (modulation circuit) 702 is monitored in comparator 703 and a correction signal generated in control logic circuit 704 is applied as input to an adder provided on the input side of the mixer such that the LO signal supplied from mixer 702 is eliminated. By means of this configuration, local leaks can be reduced even when the mixer is made up by using parts having large discrepancies in characteristics.
However, a configuration that employs the above-described first to fourth examples of the background art for correcting local leaks or DC offsets suffers from the problems described below.
The first problem is the occurrence of spike noise caused by the correction error produced when the changes in DC offset that result from frequency hopping cannot be accurately corrected. This phenomenon is explained below using FIG. 8.
FIG. 8 shows an example of a signal waveform of a reception apparatus that carries out frequency hopping by the three frequencies F1, F2, and F3.
In the output of the mixer, a waveform appears that contains an LO signal that corresponds to F1, F2, and F3, DC offset shown by a broken line, and the received signal (not shown). In FIG. 8, only changes of the DC offset are extracted and shown.
The DC offset can be corrected by generating an ideal correction value (see “ideal correction value” in FIG. 8) having an absolute value equal to the DC offset but with an inverted polarity and then by adding this ideal correction value to the output signal of the mixer. Here, generation of the ideal correction value requires both the high-speed detection of changes in the DC offset and the use of a DAC that features both high speed and high-resolution. These requirements increase the cost of the radio apparatus. When a simple DAC is used to limit an increase in cost, a correction value that can keep up with change of the DC offset cannot be generated (see “actual correction value” of FIG. 8), and a correction error is produced in time intervals (frequency transition intervals) in which the radio frequency transitions due to frequency hopping.
As previously described, the occurrence of a DC offset in the output of a mixer suppresses gain of the VGA and thus reduces the reception sensitivity, and the DC offset is therefore preferably eliminated before the baseband signal that is supplied as output from the mixer is applied to the VGA.
However, when a circuit for correcting the DC offset that uses a DAC is provided on the output side of a mixer, a correction error that is generated in the above-described frequency transition intervals is supplied to the VGA, whereby this correction error is amplified to produce the spike noise as shown in the “correction output” of FIG. 8.
The second problem is the generation of large spike noise by divergence of the correction timing during correction of the DC offset. FIG. 8 shows an example of the correction error that occurs in frequency transition intervals in frequency hopping, but a correction error also occurs when the timing of correction diverges from the timing of frequency hopping. The spike noise that occurs as a result of this divergence of the correction timing also occurs in a configuration that does not employ a DAC as in the second example of the background art. In other words, when the timing of frequency hopping diverges from the timing of switching switches 501 shown in FIG. 5, a correction error is produced and spike noise is generated.
The correction error that is caused by this divergence of correction timing is particularly conspicuous in a radio apparatus in which high-speed frequency hopping is required. For example, the completion of frequency hopping in a 9.47 ns period for each 312.5 ns is demanded in a UWB communication apparatus. This type of high-speed frequency hopping was not demanded of radio apparatuses of the prior art, but as the demand for higher speeds for data transfer rates continues to increase into the future, the potential exists that even faster frequency hopping will be required.
In radio apparatuses in recent years, the trend is toward longer wiring length due to, for example, the increasing scale of LSI that accompanies the increase in functions, and as a result, skew adjustment has become necessary for the timing of switching frequencies and the timing of the output of a DC offset correction value. These requirements increase the design man-hours of the radio apparatus and therefore increase costs.
The third problem is the generation of spike noise due to the glitch noise of the DAC in a configuration that uses a DAC to correct DC offset.
When supplying an analog value that corresponds to the digital code that is received as input, a DAC may at times instantaneously supply an analog value (glitch noise) that differs from the digital code preceding alteration or the digital code after alteration.
The glitch noise supplied from a DAC becomes a correction error of the DC offset and therefore, as in the first and second problems described above, is amplified by a succeeding VGA to result in spike noise.
Although the amelioration of this phenomenon necessitates the use of a DAC in which glitch noise does not occur, the use of a high-performance DAC in which glitch noise does not occur tends to increase the scale and power consumption of the offset correction circuit for correcting DC offset. Although a method can be considered in which the output signal of the DAC is subjected to filtering to reduce the glitch noise, in this case the design is complicated by the need to prevent deterioration of the settling characteristics of the DAC.
The fourth problem is the occurrence of spike noise resulting from the occurrence of correction error, divergence of the correction timing, or glitches of the DAC, as in the first to third problems described hereinabove, even in configurations that correct local leaks shown in the fourth example of the background art.
When spike noise is supplied as output from a mixer provided in a transmission apparatus, the operation of an amplification circuit for transmission of a succeeding stage becomes saturated, raising the danger of the generation of unwanted radiation and the failure to meet standards established by laws. Implementing skew adjustment or improving DAC characteristics to reduce this spike noise tends to increase the circuit scale or power consumption of the radio apparatus.